The present invention relates to a wireless terminal, for
example a mobile phone handset.
Wireless terminals, such as mobile phone handsets, typically
incorporate either an external antenna, such as a normal mode helix or meander line
antenna, or an internal antenna, such as a Planar Inverted-F Antenna (PIFA) or similar.
Such antennas are small (relative to a wavelength) and
therefore, owing to the fundamental limits of small antennas, narrowband. However,
cellular radio communication systems typically have a fractional bandwidth of 10%
or more. To achieve such a bandwidth from a PIFA for example requires a considerable
volume, there being a direct relationship between the bandwidth of a patch antenna
and its volume, but such a volume is not readily available with the current trends
towards small handsets. Hence, because of the limits referred to above, it is not
feasible to achieve efficient wideband radiation from small antennas in present-day
A further problem with known antenna arrangements for wireless
terminals is that they are generally unbalanced, and therefore couple strongly to
the terminal case. As a result a significant amount of radiation emanates from the
terminal itself rather than the antenna. A wireless terminal in which an antenna
feed is directly coupled to the terminal case, thereby taking advantage of this
situation, is disclosed in our co-pending unpublished International patent application
PCT/EPO1/08550 (Applicant's reference PHGB010056). When fed via an appropriate matching
network the terminal case acts as an efficient, wideband radiator.
Disclosure of Invention
An object of the present invention is to provide a compact
wireless terminal having efficient radiation properties without the need for a matching
According to the present invention there is provided a
wireless terminal comprising a ground conductor and a transceiver coupled to an
antenna feed, wherein the antenna feed is coupled directly to the ground conductor
via a capacitor formed by a conducting plate and a portion of the ground conductor
and wherein a matching slot, partially located underneath the conducting plate,
is provided in the ground conductor.
The location of a slot beneath the conducting plate performs
much of the function of a conventional matching circuit, thereby simplifying implementation
of a wireless terminal. More than one slot may be provided, and a slot may be folded
as dictated by space or other requirements.
The present invention is applicable to any wireless communication
system where the use of a large antenna is not appropriate. Since the coupling capacitor
is small, it is ideally suited to an RF IC or module, where the coupling capacitor
would be part of the module. It is particularly useful in wireless systems that
feature multiband or wideband operation.
The present invention is based upon the recognition, not
present in the prior art, that the impedances of an antenna and a wireless handset
are similar to those of an asymmetric dipole, which are separable, and on the further
recognition that the antenna impedance can be replaced with a non-radiating coupling
Brief Description of Drawings
Embodiments of the present invention will now be described,
by way of example, with reference to the accompanying drawings, wherein:
- Figure 1 shows a model of an asymmetrical dipole antenna, representing the combination
of an antenna and a wireless terminal;
- Figure 2 is a graph demonstrating the separability of the components of the
impedance of an asymmetrical dipole;
- Figure 3 is an equivalent circuit of the combination of a handset and an antenna;
- Figure 4 is an equivalent circuit of a capacitively back-coupled handset;
- Figure 5 is a perspective view of a basic capacitively back-coupled handset;
- Figure 6 is a graph of simulated return loss S11 in dB against frequency
f in MHz for the handset of Figure 5;
- Figure 7 is a Smith chart showing the simulated impedance of the handset of
Figure 5 over the frequency range 1000 to 2800MHz;
- Figure 8 is a graph showing the simulated resistance of the handset of Figure
- Figure 9 is a plan view of a single-slotted self-resonant capacitively back-coupled
- Figure 10 is a graph of simulated return loss S11 in dB against frequency
f in MHz for the handset of Figure 9;
- Figure 11 is a Smith chart showing the simulated impedance of the handset of
Figure 9 over the frequency range 800 to 3000MHz;
- Figure 12 is a plan view of a doubly-slotted self-resonant capacitively back-coupled
- Figure 13 is a graph of simulated return loss S11 in dB against frequency
f in MHz for the handset of Figure 12;
- Figure 14 is a Smith chart showing the simulated impedance of the handset of
Figure 12, over the frequency range 800 to 3000MHz;
- Figure 15 is a graph of simulated return loss S11 in dB against frequency
f in MHz for the handset of Figure 12 fed via a matching network; and
- Figure 16 is a Smith chart showing the simulated impedance of the handset of
Figure 12 fed via a matching network, over the frequency range 800 to 3000MHz.
In the drawings the same reference numerals have been used
to indicate corresponding features.
Modes for Carrying Out the Invention
Figure 1 shows a model of the impedance seen by a transceiver,
in transmit mode, in a wireless handset at its antenna feed point. The impedance
is modelled as an asymmetrical dipole, where the first arm 102 represents the impedance
of the antenna and the second arm 104 the impedance of the handset, both arms being
driven by a source 106. As shown in the figure, the impedance of such an arrangement
is substantially equivalent to the sum of the impedance of each arm 102,104 driven
separately against a virtual ground 108. The model could equally well be used for
reception by replacing the source 106 by an impedance representing that of the transceiver,
although this is rather more difficult to simulate.
The validity of this model was checked by simulations using
the well-known NEC (Numerical Electromagnetics Code) with the first arm 102 having
a length of 40mm and a diameter of 1 mm and the second arm 104 having a length of
80mm and a diameter of 1 mm. Figure 2 shows the results for the real and imaginary
parts of the impedance (R+jX) of the combined arrangement (Ref R and Ref X) together
with results obtained by simulating the impedances separately and summing the result.
It can be seen that the results of the simulations are quite close. The only significant
deviation is in the region of half-wave resonance, when the impedance is difficult
to simulate accurately.
An equivalent circuit for the combination of an antenna
and a handset, as seen from the antenna feed point, is shown in Figure 3. R1
and jX1 represent the impedance of the antenna, while R2 and
jX2 represent the impedance of the handset. From this equivalent circuit
it can be deduced that the ratio of power radiated by the antenna, P1,
and the handset, P2, is given by
If the size of the antenna is reduced, its radiation resistance
R1 will also reduce. If the antenna becomes infinitesimally small its
radiation resistance R1 will fall to zero and all of the radiation will
come from the handset. This situation can be made beneficial if the handset impedance
is suitable for the source 106 driving it and if the capacitive reactance of the
infinitesimal antenna can be minimised by increasing the capacitive back-coupling
to the handset
With these modfications, the equivalent circuit is modified
to that shown in Figure 4. The antenna has therefore been replaced with a physically
very small back-coupling capacitor, designed to have a large capacitance for maximum
coupling and minimum reactance. The residual reactance of the back-coupling capacitor
can be tuned out with a simple matching circuit. By correct design of the handset,
the resulting bandwidth can be much greater than with a conventional antenna and
handset combination, because the handset acts as a low Q radiating element (simulations
show that a typical Q is around 1), whereas conventional antennas typically have
a Q of around 50.
A basic embodiment of a capacitively back-coupled handset
is shown in Figure 5. A handset 502 has dimensions of 10×40×100mm, typical
of modem cellular handsets. A parallel plate capacitor 504, having dimensions 2×10×10mm,
is formed by mounting a 10×10mm plate 506 2mm above the top edge 508 of the
handset 502, in the position normally occupied by a much larger antenna. The resultant
capacitance is about 0.5pF, representing a compromise between capacitance (which
would be increased by reducing the separation of the handset 502 and plate 506)
and coupling effectiveness (which depends on the separation of the handset 502 and
plate 506). The capacitor is fed via a support 510, which is insulated from the
handset case 502.
The return loss S11 of this embodiment after
matching was simulated using the High Frequency Structure Simulator (HFSS), available
from Ansoft Corporation, with the results shown in Figure 6 for frequencies f between
1000 and 2800MHz. A conventional two inductor "L" network was used to match at 1900MHz.
The resultant bandwidth at 7dB return loss (corresponding to approximately 90% of
input power radiated) is approximately 60MHz, or 3%, which is useful but not as
large as was required. A Smith chart illustrating the simulated impedance of this
embodiment over the same frequency range is shown in Figure 7.
The low bandwidth is because the combination of the handset
502 and capacitor 504 present an impedance of approximately 3-j90&OHgr; at 1900MHz.
Figure 8 shows the resistance variation, over the same frequency range as before,
simulated using HFSS. This can be improved by redesigning the case to increase the
resistance, for example by the use of a slot or a narrower handset, as discussed
in our co-pending unpublished International patent application PCT/EPO1/08550.
The handset of Figure 5 requires matching to obtain reasonable
performance. There are significant advantages to being able to eliminate the need
for matching. A plan view of a modfied single band configuration which requires
no matching is shown in Figure 9. This embodiment differs from that of Figure 5
in that the 10mm square plate 506 is located 2mm above the back of the handset 502,
and in that a slot 912 of length 30mm and width 1 mm is cut in the conducting material
2mm from the edge of the handset case. The slot 912 extends under the conducting
plate 506 (as shown by dashed lines in Figure 9). The slot 912 is resonant at odd
multiples of a quarter wavelength, i.e. at &lgr;/4, 3 &lgr;/4, etc.
The slot presents a high impedance to the coupling capacitor,
thereby enabling a good match to 50&OHgr;. It is believed that the capacitor excites
a transmission line mode in the slot 912 that acts as a shunt inductance at the
antenna feed, which acts to match the response.
In the illustrated embodiment the slot 912 is located close
to the edge of the handset case 502 in order to minimise the space used, although
the slot could equally well be located on the other side of the coupling capacitor
504. Similarly, the coupling capacitor could be implemented in other positions on
the handset 502 and the slot 912 could have a range of configurations, for example
vertical, horizontal or meandering.
The return loss S11 of this embodiment, without
matching, was simulated using HFSS, with the results shown in Figure 10 for frequencies
f between 800 and 3000MHz. The resultant bandwidth at 7dB return loss is approximately
90MHz, or 4.3%. Although the bandwidth could be improved with matching, it is useful
to be able to avoid having to include matching and the bandwidth is already more
than sufficient for a Bluetooth embodiment, for example.
A Smith chart illustrating the simulated impedance of this
embodiment over the same frequency range is shown in Figure 11. This shows that
the configuration of Figure 9 also has the useful property that resonance (zero
reactance) is achieved twice, with the higher frequency resonance having the higher
resistance. This is particularly convenient, since the receive band is usually at
a higher frequency in a frequency duplex system.
A preferred transceiver architecture is to maintain a low
impedance path between the (generally low impedance) transmitter and the antenna,
and a high impedance path between the antenna and the (generally high impedance)
receiver. However, for simplicity of design it is conventional to use a 50&OHgr;
system impedance with additional matching at the transmitter and receiver as required.
This matching is lossy, and may also reduce the bandwidth seen at both the transmitter
and receiver. Hence, the removal of the need for matching is a significant advantage
of the present invention.
A dual band embodiment of the present invention is shown
in plan view in Figure 12. In this embodiment the plate 506 and slot 912 have been
moved to the top centre of the back surface of the handset 502, and a further slot
1214 has been added. The further slot 1214 is longer than the first slot 912, having
a a total length of approximately 73mm and a width of 1mm, and folded to reduce
the area it occupies.
The return loss S11 of this embodiment, without
matching, was simulated using HFSS, with the results shown in Figure 13 for frequencies
f between 800 and 3000MHz. It can clearly be seen that this design allows dual,
tri or multiband operation. The slots 912, 1214 are resonant at odd multiples of
&lgr;/4, and can therefore be arranged to give individual or combined resonances.
The first resonance (at approximately 1GHz) is the &lgr;/4 resonance of the longer
slot 1214. The second resonance (at approximately 1.8GHz) is the &lgr;/4 resonance
of the shorter slot 912. The third resonance (at approximately 2.8GHz) is the 3
&lgr;/4 resonance of the longer slot 1214. It is clear, for example, that, with
some modification, this configuration can be used for GSM, DCS1800 and Bluetooth.
The resultant bandwidths at 7dB return loss for the three
resonances are approximately 15MHz (1.5%), 110MHz (5.9%) and 110MHz (3.9%). The
bandwidth of the 1 GHz resonance is small, but the other bandwidths are good. A
Smith chart illustrating the simulated impedance of this embodiment over the same
frequency range is shown in Figure 13. The rapid changes in impedance in the Smith
chart reflect the narrow-band nature of the first resonance.
The self-resonance of each slot 912,1214 is independently
variable via its position under the feeding capacitor 504: as the slot 912,1214
is progressively moved under the plate 506 the effect of its nominal shunt inductance
increases. Also, each slot 912,1214 is high impedance at its open end and low impedance
at its shorted end. Hence, the resistance could be varied by tapping off at various
points along the slot. The capacitor can also be made asymmetric to allow for such
tapping to be performed, to some extent
Embodiments of the present invention may also be used in
conjunction with matching. As an example, simulations of the dual slot configuration
illustrated in Figure 12 in conjunction with a simple "L" matching circuit similar
to that used for the basic embodiment of Figure 5 were performed. Results for the
return loss S11 are shown in Figure 15 for frequencies f between 800
and 3000MHz. It can be seen that a very wide bandwidth is achieved (a 3dB bandwidth
of approximately 1.4GHz). This could be enhanced further with a more elaborate matching
circuit. A Smith chart illustrating the simulated impedance of this embodiment over
the same frequency range is shown in Figure 16.
In the above embodiments a conducting handset case has
been the radiating element. However, other ground conductors in a wireless terminal
could perform a similar function. Examples include conductors used for EMC shielding
and an area of Printed Circuit Board (PCB) metallisation, for example a ground plane.
From reading the present disclosure, other modifications
will be apparent to persons skilled in the art. Such modifications may involve other
features which are already known in the design, manufacture and use of wireless
terminals and component parts thereof, and which may be used instead of or in addition
to features already described herein.