The invention relates in general to modulators. In particular
the invention relates to a modulator structure which is suitable for dual-band or
triple-band mobile stations.
In modulation a carrier frequency is modified in a certain
way so that the data to be transmitted using the radio signal is carried, for example,
in the changes of the phase or amplitude of the carrier frequency. There are many
modulation methods, which differ in the sense which properties of the carrier wave
are modulated and how they are modulated. The arrangement that performs the modification
of the carrier wave is called a modulator. There are also many types of modulators.
A direct conversion modulator, for example, may be used in mobile stations. In a
direct conversion modulator the modulation is performed directly in the carrier
frequency; there is no intermediate frequency in the modulation process.
The most important characteristics of a modulator are the
linearity and the signal-to-noise ratio S/N. The signal-to-noise ratio comprises
the ratio of the transmit signal to the background noise at the transmit frequency
range (TX), and the ratio of the transmit signal to the noise generated at the receive
frequency range (RX). It is possible to enhance the signal-to-noise ratio at the
RX frequency range by filtering the modulated signal using a suitable filter. The
enhancement of the signal-to-noise ratio at the TX frequency range is not feasible
after the modulator.
Usually to filtering at the RX frequency range is done
using a low pass filter, most preferably a duplex filter after the power amplification.
Table 1 presents the signal-to-noise ratio limits for the receive frequency band
defined for the GSM (Global System for Mobile Communications), PCN (Personal Communication
Network) and PCS (Personal Communication System) networks.
In the table N 100 kHz defines the noise level generated
by the transmitter chain in the reception at a frequency band with the width of
100 kHz. The noise is measured at a power level where the noise level is at a maximum.
This power level is in practice the maximum transmit power. The measured noise power
level is normalized from the 100 kHz width to a width of 1 Hz, whereby the level
decreases by 50 dB and is in the column N 1 Hz. Thus the measurement is performed
at a wide bandwidth, but it is normalized to a narrower band by calculation. There
are also individual noise level differences between similar transmitters, and therefore
there is a margin of 3 dBm, which is subtracted from the above mentioned level limits.
Pout is the output power of the transmitter. S/N is the signal-to-noise
ratio (dBc, decibel carrier) generated at the carrier frequency, which is determined
by the distance between the maximum output power and the normalized noise, to which
the margin is added.
Table 1 The signal-to-noise ratio limits for the receive frequency band
N 1 Hz
It is also possible to filter the noise by placing filters
between the modulator of the transmit chain and the receiver.
Figure 1 shows the basic effect of a low-pass filter on
radio signals. The original transmit signal 1 is mirrored into the frequency domain
over the modulated center frequency. Figure 1A shows that the final result 2 of
the modulated but unfiltered transmit signal extends both into the intended transmit
range TX and partly into the receive range RX. Figure 1 B shows that the final result
4 of the modulated transmit signal 1, which is filtered with the filter function
3, extends mainly into the intended transmit range TX, and only slightly into the
receive range RX.
Previously there is also known a so called Gilbert cell,
which is generally used in integrated multiplicator circuits of telecommunication
systems, particularly in mobile stations. Multiplicator circuits are used for instance
in integrated RF (Radio Frequency) and intermediate frequency sections, such as
in the modulator, the mixer and the regulated amplifier.
US patent US 5,172,079
describes a modulator using a Gilbert cell with current mirrors in its
A prior art multiplicator, such as a squaring multiplicator,
is based on a mathematical formula where the difference of the squares of the sum
and difference of two signals produces the product of the signals:
where x is the first signal and y is the second signal.
The squaring is performed for instance by a MOS (Metal
Oxide Semiconductor) transistor where the drain current is proportional to the square
of the gate-source voltage. A multiplicator can also be realized with bipolar transistors.
The collector current of a bipolar transistor iC is:
where the shown parameters are the bipolar transistors saturation current IS,
the base-emitter voltage VBE, and the thermal voltage VT.
The exponent function is here approximated by the first four terms of an infinite
exponential Taylor series:
and the shown parameters are the bipolar transistor's base-emitter voltage vBE
and the thermal voltage VT.
Mixers use generally a double balanced structure where
the outputs of four differently phased mixers are added in order to equalize the
harmonic interference. The balancing is presented mathematically as follows:
When the first four terms of the exponential Taylor series
are substituted in the formula we still have the product of the input signals:
When we substitute the terms x and y for the terms representing
the output signals the final mathematical formula can be written in the form:
Figure 2 shows a Gilbert cell known per se, which
is used for instance to realize integrated RF and intermediate frequency sections,
such as a regulated amplifier and a mixer. In a Gilbert cell two input voltages
recur as one output voltage, in other words the voltage difference at the outputs
is the product of the voltage differences at the inputs. The first voltage difference
is connected to the terminals VX+ and VX-, from which the
voltages are supplied to the bases of the transistors Q1 and Q2 and correspondingly
to the bases of the transistors Q4 and Q5. The other voltage difference is connected
to the terminals VY+ and VY-, from which the voltage is amplified
by the transistors Q3 and Q6. The transistors Q3 and Q6 are connected via the resistors
RE1 and RE2 to the field effect transistor (FET) Q7 which
is controlled by the biasing voltage VBIAS and connected to the negative
operating voltage. The transistors Q1 and Q5 amplify the positive voltage difference
VX+ and VX which is connected to the outputs VOUT+
and VOUT-. The above mentioned circuit is connected to the positive operating
voltage via the resistors RL1 and RL2. The transistors Q2
and Q4 amplify the negative voltage difference VX+ and VX-
which is connected to the outputs VOUT+ and VOUT-.
Figure 3 shows the circuit arrangement of the switch transistors
in a direct conversion modulator. The arrangement comprising the switch transistors
is often called the switching arrangement or switching block of a modulator. Noise
due to the local oscillator can be to a large extent eliminated by using balanced
switching arrangement, for example pairs of transistors that have very similar characteristics.
Figure 3 shows the switch transistors Q8, Q9, Q10, Q11,
Q12, Q13, Q14 and Q15 of a direct conversion modulator. These transistors form four
balanced transistor pairs. Four differently phased signals LO0, LO180,
LO90 and LO270 from the local oscillator are supplied as control
signals to the transistors. The transmit signals TXI0 and TXI180
with opposite phases and their complements TXQ0 and TXQ180
are supplied to the emitters of the transistors. The output signals OUT+ and OUT-
are obtained at the transistor collectors.
A problem in known devices is that a mobile station operating
at two or three frequency bands, i.e. a dual-band or triple-band mobile station,
requires many filters, which occupy too much space in a mobile station. In practice
the realization of a small-sized RF section in this way is impracticable, and therefore
there is a tendency to avoid filtering. However, when filtering is omitted the modulator
must be of a particularly high quality.
A further problem in known devices is that they require
separate filters between the modulator and the transmitter.
Against this background, the invention resides in minimizing
in the modulator the noise accumulated to the modulated signal.
A modulator according to the invention is claimed in claim
Accordingly, the present invention provides a modulator
structure having good signal-to-noise properties. Furthermore, the invention provides
a compact modulator structure that can be used in many frequency bands.
A mobile station according to a further aspect of the invention
is claimed in claim 7.
The noise that a modulator causes to the receiver frequency
range comprises the noise of the switch transistors in the modulator, the noise
of the local oscillator, and the noise of the signal carrying the information to
be transmitted. In a balanced modulator structure the noise due to the switch transistors
is dampened so that it is not the dominating component in noise. Current state-of-the-art
bipolar transistors nowadays have usually low enough noise level.
The noise in the information signal can be a dominating
factor, especially in direct conversion modulators. Therefore it is advantageous
to filter the possible noise component away from the signal carrying the information
to be transmitted. In a modulator according to the invention, this filtering is
carried out in the modulator.
A modulator here refers to an arrangement where information
signal, i.e. the signal carrying the data to be transmitted, and the signal having
the carrier frequency interact. The information signal is usually coupled to the
driver arrangement of a modulator and the signal from the local oscillator to the
switching arrangement. The driver and switching arrangements are coupled to each
other in such a way that the information signal modifies the carrier wave from the
local oscillator. In a modulator according to the invention, a low-pass filter for
attenuating high frequency interference is integrated to the driver arrangement
of the modulator.
In one embodiment the modulator is of the direct conversion
In one embodiment the low-pass filter is located in the
current mirror of the multiplicator circuit of the modulator.
In one embodiment the low-pass filter is located between
the transistors of the transistor pair which mirrors the current in the current
mirror, whereby the low-pass filter filters the signal path between the transistor
supplying the control and the transistor generating the mirrored current.
In one embodiment the low-pass filter is formed by an RC
low-pass arrangement where the resistor is connected in series with the signal path
and the capacitor is connected in parallel regarding the signal's propagation direction
behind the resistor, to the common potential of the transistor pair.
In one embodiment the common potential of the transistor
pair is connected to the negative potential of the operating voltage.
The invention relates also to a mobile station having a
low-pass filter in the transmitter chain for attenuating high frequency interference.
Advantageously, the low-pass filter is located in the direct conversion modulator
of the transmitter.
An advantage of the invention is that the required low-pass
filter can be integrated into the modulator. The low-pass filter dampens the noise
due to the information signal before the information signal is mixed with the carrier
wave. Therefore the low-pass filter integrated to the modulator can replace the
filters that are currently placed after the power amplification of the modulated
signal and that are chosen separately for each frequency band.
A further advantage of the invention is that with a low-pass
filter according to the invention the transmission chain has a better signal-to-noise
ratio than previous solutions.
The invention is described in detail below with reference
to the enclosed drawing, in which
- Figure 1
- shows the basic effect of a low-pass filter at the frequency ranges of a transmitter
and a receiver;
- Figure 2
- shows a Gilbert cell known per se;
- Figure 3
- shows the circuit arrangement of the switch transistors in a direct conversion
- Figure 4
- shows a schematic drawing of a modulator according to a preferred embodiment
of the invention;
- Figure 5
- shows elements of the preferred embodiment in a circuit diagram of a direct
conversion modulator according to the preferred embodiment of the invention;
- Figure 6
- shows the current mirror of a direct conversion modulator according to the preferred
embodiment of the invention; and
- Figure 7
- shows in a block diagram some elements of a mobile station according to a preferred
form of the invention.
Figures 1, 2 and 3 are discussed in the section concerning
Figure 4 shows a schematic drawing of a modulator 400 according
to a preferred embodiment the invention. The modulator 400 comprises a switching
arrangement 410 to which the local oscillator is coupled (arrow 441 in Figure 4).
It further comprises a driver arrangement 420 to which the information signal is
couples (arrow 442 in Figure 4). The modulated signal, i.e. the carrier wave that
has been modified according to the information signal, is derived from the modulator
In Figure 4 the signal from the local oscillator, the information
signal and the modulate signal are presented with single arrows. In actual implementation,
for example, the signal from the local oscillator is usually coupled to the modulator
in different phases. Similarly, the information signal may be coupled to the modulator
using more than one input points, and the modulated signal may be obtained from
the modulator via many output points. The driver and switching arrangements may
also be coupled to each other in many points. The line in Figure 4 in between the
arrangements 410 and 420 only schematically indicates that the blocks are coupled
to each other. It does not implicate that they should be coupled to each other in
only one point.
The driver arrangement 420 comprises certain driver components
and, in addition to these components, in a modulator 400 according to the invention,
it comprises a filter arrangement 430. This low-pass filter arrangement is used
to eliminate the noise in the information signal to a large extent. Preferably the
filter arrangement is such that is it possible to construct it as a part of the
Figure 5 shows on two pages, in the figures 5A and 5B,
the elements of the invention in a circuit diagram of a direct conversion modulator
according to a preferred form of invention. In figure 5A we see how the transmit
signals TXI0 and TXI180 with opposite phases and their complements
TXQ0 and TXQ180 through the biasing current IBIAS
control the connection between the positive operating voltage potential V+ and the
current mirrors 7 of figure 5B. The current IBIAS controls the currents
with the aid of the current mirror 5 comprising the arrangement of the transistors
Q16, Q17 and Q18. The currents are further supplied to the current mirrors 7 through
the resistors R1, R2, R3 and R4, and through the control arrangement 6 comprising
the transistors Q19, Q20, Q21 and Q22 which are controlled by transmit signals.
Thus the transmit signals TXI0, TXI180, TXQ0, TXQ180
and the biasing current IBIAS control the modulator with the aid of the
current mirrors 7. The transistors in the current mirrors 7 in Figure 5B present
an example of driver components in a driver arrangement 420 in Figure 4.
Due to the low operating voltage and the required high
output power we can see in the modulator an arrangement having only two transistors
in series between the output OUT+, OUT- and the negative operating voltage potential
V-. These two transistors are the switch transistors 8 and the transistors of the
current mirror 7.
The RC low-pass filters (RC, resistor-capacitor) according
to the invention which filter the high frequency noise of the transmit signals TXI0,
TXI180, TXQ0, TXQ180 and the biasing current IBIAS
are seen in the current mirrors 7 formed by the resistors R5, R6, R7, R8 and the
capacitors C1, C2, C3, C4 between the transistors Q23, Q24, Q25, Q26, Q27, Q28,
Q29, Q30. The capacitors C1, C2, C3, C4 are connected to the negative potential
V- of the operating voltage, or to any potential having a sufficiently low noise.
Regarding the signal passage the filter is located as late as possible to enable
a high signal-to-noise ratio. Otherwise the unfiltered noise would be mixed to the
reception frequency range and the signal-to-noise ratio would decrease.
Figure 6 shows a more unconstrained arrangement of one
current mirror located in the current mirrors 7 of figure 4B. On the signal path
behind the controlling transistor Q23 there is an RC filter 9 comprising a series
resistance R5 and a parallel capacitance C1. The signal is further supplied to the
transistor Q24 generating the mirrored current. In Figure 6 the resistor R5 and
the capasitor C1 (and how they are connected to the current mirror) present an example
of the filter arrangement 430 in Figure 4.
Figure 7 shows some parts of a mobile station utilizing
the invention. The mobile station comprises according to prior art a processor 11,
a memory 12, a display 13, a keyboard 14, and an audio block 15 with a microphone
16A and a speaker 16B. The modulator according to the invention, which contains
a low-pass filter, is located in the transceiver block 17, which is connected to
the antenna 18.
The invention is not limited to concern only the presented
embodiment examples, but many modifications are possible within the inventive idea
defined by the claims. A mobile station according to the invention may be a mobile
station related to any cellular system.